High speed latch circuits using gated diodes

ABSTRACT

A sense amplifier circuit comprises (1) an isolation device comprising a control terminal and first and second terminals, the first terminal of the isolation device coupled to a signal line, (2) a gated diode comprising first and second terminals, the first terminal of the gated diode coupled to the second terminal of the isolation device, and the second terminal of the gated diode coupled to a set line; and (3) control circuitry coupled to the control terminal of the isolation device and adapted to control voltage on the control terminal of the isolation device in order to enable and disable the isolation device. A latch circuit further comprises a precharge device comprising a control terminal and first and second terminals, the first terminal of the precharge device coupled to a power supply voltage, and the second terminal of the precharge device coupled to the first terminal of the isolation device.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.10/933,706, U.S. Pat. No. 7,116,594, filed Sep. 3, 2004, which isincorporated by reference herein. This application is related to anapplication, the disclosure of which is hereby incorporated byreference, by W. Luk and R. Dennard, entitled “Amplifiers Using GatedDiodes,” U.S. Pat. Ser. No. 10/751,714, U.S. patent applicationPublication No. 2005/0145895, filed on Jan. 5, 2004, and assigned toInternational Business Machines, Inc.

FIELD OF THE INVENTION

The present invention relates to semiconductors and, more particularly,relates to semiconductor devices and circuits using the same foramplifying signals.

BACKGROUND OF THE INVENTION

In the design of integrated circuits, the ability to detect smallchanges in voltage or current allows for realization of both highperformance and low power consumption. This is possible becauseinformation indicating state of a signal can be detected and passed tosubsequent stages of a circuit without having to wait for the signal toswing through its entire range, resulting in circuits with faster speedand lower power. Such technology is commonly used in memory arrays,which allow for high-speed access of individual memory elements. Thistechnology can also be used to improve performance and power consumptionwhen driving long wires and large capacitive loads, as well as forinterfacing low voltage logic with regular logic operating under fullsupply voltage (e.g., Vdd). The enabling circuit for this technologytypically is a sense amplifier circuit, which converts a small signalchange (from the output of a low voltage circuit or from a signalsource) into a relatively large signal that can be interfaced with therest of the circuit.

In conventional single-ended, small signal sense amplifier circuits suchas “class A” sense amplifier circuits, there are a number of items thatare very difficult to control: biasing of the operating point; stabilityof the reference voltage; biasing current; sensitivity to thresholdvoltage; and process and temperature variations. This is especially truefor circuits using future technology, due to increasing high leakagecurrent and low supply voltage as transistors are scaled smaller, makingsuch circuits very sensitive to voltage, temperature and processvariations. For conventional differential-sense circuits, due to theincreasing statistical variation between adjacent transistors in futuretechnology, the advantage of differential mode small signal sensing isdiminishing.

Another widely used circuit is a latch circuit. Latch circuits are usedto hold data and logic states in large-scale integrated circuits. In apipelined architecture, synchronous data flow is governed by a referenceclock signal, which controls individual latches and latch-basedregisters, which are circuit blocks that either hold data or allow it topass into the next pipeline stage. This technology significantlyincreases data throughput, which allows for high performance logic andmemory circuits. By combining sense amplifier circuits and latchfunctionality into a single circuit block, high bandwidth signalamplification can be achieved.

Existing circuit techniques involve feeding the output of a senseamplifier circuit into a latch, which incurs the delay of two separatestages, thus making it difficult to attain high speed operation. Thus,there is a need to provide improved sense amplifier circuits, includingcircuits latching data, for uses such as signal sensing.

SUMMARY OF THE INVENTION

The present invention provides sense amplifier and latch circuits usinggated diodes. Illustratively, the present invention presents new classesof sense amplifier and latch circuits based on gated diodes. The senseamplifier and latch circuits disclosed herein can used digital control,can detect and amplify small signals, and can function properly androbustly under a wide range of operating conditions of supply voltage,temperature, and process variation.

In a first aspect of the invention, a sense amplifier circuit isdisclosed that comprises an isolation device comprising a controlterminal and first and second terminals, the first terminal of theisolation device coupled to a signal line. The sense amplifier circuitalso comprises a gated diode comprising first and second terminals, thefirst terminal of the gated diode coupled to the second terminal of theisolation device, and the second terminal of the gated diode coupled toa set line. The sense amplifier circuit additionally comprises controlcircuitry coupled to the control terminal of the isolation device andadapted to control voltage on the control terminal of the isolationdevice in order to enable and disable the isolation device. The controlcircuitry is additionally coupled to the set line and adapted to controla voltage on the set line. The signal line is adapted to be coupled toan input signal, and the second terminal of the isolation device may beused to derive an output for the sense amplifier circuit.

The sense amplifier circuit may also comprise an output devicecomprising an input and an output, the input of the output devicecoupled to the first terminal of the gated diode and to the secondterminal of the isolation device. The output of the output device isadapted to be the output of the sense amplifier circuit. The outputdevice is further adapted to produce an output signal on the output ofthe sense amplifier circuit based on a voltage on the first terminal ofthe gated diode.

In a second aspect of the invention, a latch circuit is disclosed thatcomprises a pass device comprising a control terminal and first andsecond terminals, the first terminal of the pass device coupled to asignal line, the control terminal of the pass device coupled to a firstclock line. A precharge device in the latch circuit comprises a controlterminal and first and second terminals, the control terminal of theprecharge device coupled to a second clock line, the first terminal ofthe precharge device coupled to a power supply voltage, and the secondterminal of the precharge device coupled to the first terminal of thepass device. The latch circuit further comprises a gated diodecomprising first and second terminals, the first terminal of the gateddiode coupled to the second terminal of the pass device, and the secondterminal of the gated diode coupled to a third clock line. An outputdevice, as part of the latch circuit, comprise an input and an output,the input of the output device coupled to the first terminal of thegated diode and to the second terminal of the pass device, the output ofthe output device adapted to be the output of the latch circuit, theoutput device adapted to produce an output signal on the output of thesense amplifier circuit based on a voltage on the first terminal of thegated diode. The signal line is adapted to be coupled to an inputsignal.

A more complete understanding of the present invention, as well asfurther features and advantages of the present invention, will beobtained by reference to the following detailed description anddrawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A shows exemplary symbols used for a first n-type gated diode;

FIG. 1B shows an example of a side view of the first n-type gated diodeformed in a semiconductor;

FIG. 1C shows an exemplary representative circuit used for modeling thefirst n-type gated diode shown in FIG. 1B;

FIG. 1D shows an example of a side view of the first n-type gated diodeformed in Silicon-On-Insulator (SOI);

FIG. 1E shows an exemplary representative circuit used for modeling thefirst n-type gated diode shown in FIG. 1D;

FIG. 2A shows exemplary symbols used for a second n-type gated diode;

FIG. 2B shows an example of a side view of the second n-type gated diodeformed in a semiconductor;

FIG. 3 is a graph illustrating typical capacitance of the gatecapacitance Cgs (obtained by the derivative of charge with respect tovoltage, dq/dv) versus voltage between the gate and source (Vgs) for ann-type gated diode in bulk silicon, for a number of different gateareas;

FIG. 4A shows an example of a voltage boosting circuit using acapacitor;

FIG. 4B shows graphs illustrating gain for the voltage boosting circuitof FIG. 4A;

FIG. 5A shows an example of a gated diode voltage boosting circuit usedfor an amplifier;

FIG. 5B shows an exemplary representative circuit for the gated diodeamplifier of FIG. 5A when the gated diode is turned OFF;

FIG. 5C shows an exemplary representative circuit for the gated diodeamplifier of FIG. 5A when the gated diode is turned ON;

FIG. 6 shows graphs illustrating gain for a gated diode amplifier when agated diode is used as the charge storage and transfer device;

FIG. 7 shows an example of a top view of two of the first n-type gateddiode formed in a semiconductor;

FIG. 8 is an example of a gated diode sense amplifier circuit using afixed control voltage on an isolation device;

FIG. 9 shows a number of waveforms for the sense amplifier circuit ofFIG. 8;

FIG. 10 is an example of a sense amplifier circuit using a gated diodeamplifier and control circuitry adapted to control certain elements ofthe amplifier circuit;

FIG. 11 is another example of a sense amplifier circuit, interfacingwith an optional keeper circuit, using a gated diode amplifier andcontrol circuitry adapted to control certain elements of the amplifiercircuit;

FIGS. 12 and 13 are additional examples of sense amplifier circuits,each of which interfaces with an optional keeper, and uses a gated diodeamplifier and control circuitry adapted to control certain elements ofthe amplifier circuit;

FIG. 14 illustrates an example of a sense amplifier circuit interfacingwith a memory array;

FIGS. 15A and 15B are graphs illustrating waveforms for an exemplarygated diode sense amplifier circuit shown in FIG. 10;

FIGS. 16A and 16B are graphs illustrating waveforms for anotherexemplary gated diode sense amplifier circuit shown in FIG. 13;

FIGS. 17A, 17B, and 17C are graphs illustrating waveforms for a memorycell for a memory array and for operation of two types of senseamplifier circuits in the memory array;

FIG. 18 is an example of a sense amplifier latch circuit;

FIG. 19 depicts exemplary voltage waveforms for the sense amplifierlatch circuit of FIG. 18;

FIGS. 20–23 are additional examples of sense amplifier latch circuits;

FIG. 24 is a side view of a typical Metal-Oxide-Semiconductor FieldEffect Transistor (MOSFET) including source/drain extensions and halos;

FIG. 25 is a graph of capacitance versus gate voltage for the MOSFET ofFIG. 24; and

FIG. 26 is a graph of capacitance versus gate voltage for a MOSFETwithout source/drain extensions and halos.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

The present invention provides improved sense amplifier circuits andimproved latch circuits. For ease of reference, the following disclosureis separated into an Introduction section, a Sense Amplifier CircuitsUsing Gated Diodes section, a Sense Amplifier Latch Circuits section andan Improved Gated Diode Structure for Low Vt, Low Vt Fluctuation and LowParasitic Capacitance section.

Introduction

A patent application by inventors W. Luk and R. Dennard, entitled“Amplifiers Using Gated Diodes,” U.S. patent Ser. No. 10/751,714, filedon Jan. 5, 2004, the disclosure of which is hereby incorporated byreference, discloses amplifier circuits using gated diodes. The present“Introduction” section presents information related to using gateddiodes in amplifier circuits. Additional information is presented in the“Amplifiers Using Gated Diodes” patent application.

The term “gated diode” as used herein refers to a two terminalsemiconductor device comprised of a source (one terminal) and a gate(another terminal), where a relatively large amount of charge is storedin an inversion layer when the gate to source voltage (Vgs) is above(for an n-type gated diode) a threshold voltage, and substantially smallamount, orders of magnitude smaller, or no charge is stored otherwise.As a result, the equivalent capacitance of the two terminalsemiconductor device is nonlinear, meaning that the two terminalsemiconductor device has a large capacitance when the voltage on thefirst terminal relative to the second terminal is above the thresholdvoltage and has a very small capacitance when the voltage on the firstterminal relative to the second terminal is below the threshold voltage.A gated diode is an example of a two terminal semiconductor device. Anytwo terminal semiconductor device may be used comprising the propertythat the two terminal semiconductor device has a large capacitance whena voltage on the first terminal relative to the second terminal islarger than a predetermined voltage by typically a slight amount, and asubstantially small capacitance, orders of magnitude smaller, when thevoltage on the first terminal relative to the second terminal is lessthan the predetermined voltage. The predetermined voltage is called athreshold voltage herein. For instance, for a gated diode created usingn-type Field Effect Transistor (FET) technology, voltages above athreshold voltage cause a large amount of charge to be stored in aninversion layer and voltages below the threshold voltage cause asubstantially smaller amount of charge, orders of magnitude smaller, orno charge to be stored.

As shown in the following figures, in a conventional FET setting, agated diode can be formed by the source and the gate of a three terminalFET device (either n-type or p-type), with the drain floating (e.g.,disconnected or nonexistent), as shown in FIGS. 1A, 1B, and 1D.Sometimes the source and drain of such a FET can be connected togetherat the same potential and may be viewed as two gated diodes connected inparallel, as shown in FIGS. 2A and 2B. In this disclosure, these twosituations are used interchangeably. And without specifying explicitly,a gated diode is referred to as just the first basic form, only a sourceand a gate of a semiconductor device.

FIG. 1A shows exemplary symbols used for a first n-type gated diode.Symbol 190 is an exemplary symbol for a first n-type gated diode shownin FIGS. 1A and 1B. FIG. 1B shows an example of a side viewcross-section of the first n-type gated diode 100 formed in asemiconductor. The first n-type gated diode 100 comprises a gateinsulator 120 formed between a gate 115 (e.g., N+ doped polysilicon) anda p-well 130, a source diffusion region 110, two Shallow TrenchIsolation (STI) regions 105 and 125, an optional n isolation band 140,and a p-substrate 135. As described below, the dopant concentration inp-well 130 substantially controls the threshold voltage of the gateddiode 100. An inversion layer 126 is formed when the threshold voltage,Vt, is reached on the gate to source voltage, Vgs.

In FIG. 1B, capacitance components exist between the gate 115, source110, body (e.g., the volume of the p-well 130 under the gate 115 andbetween the source 110 and the STI region 125), and substrate 135. Forexample, four capacitances may be derived. These capacitances are calledCg_gd(ON), Cg_gd(OFF) for the gated diode, and CL(ON), CL(OFF) for theload. Although additional capacitance components may be used duringmodeling, the Cg_gd(ON), Cg_gd(OFF), CL(ON), and CL(OFF) capacitancesare considered to be sufficient for modeling amplification by a gateddiode. The ON and OFF gated diode capacitances, Cg_gd(ON) andCg_gd(OFF), respectively, in terms of internal capacitances are shown inFIG. 1C.

FIG. 1C shows an exemplary equivalent circuit used for modeling thefirst n-type gated diode shown in FIG. 1B. In FIG. 1C (with appropriatereference to FIG. 1B), “R” is a resistance representing the ON/OFFinversion channel of the gated diode, Cov is an overlap capacitance fromthe gate 115 to the source 110, Csb is capacitance from the source 110to the body substrate which is typically grounded for bulk silicon, Coxis the capacitance of the gate insulator 120 (typically an oxide ordielectric), Cgb is the capacitance between the gate 115 and the bodysubstrate, and CL is the capacitance of the external load.

The equivalent ON and OFF load capacitances to ground, CL(ON) andCL(OFF), respectively, include the external load CL. The internalcapacitances of the gated diode are also shown. Exemplary equationssuitable for modeling and that use the representative circuit of FIG. 1Care as follows, where “ON” indicates that the gated diode is turned onand “OFF” indicates that the gated diode is turned off:

R(ON)=small;

Cgd(ON)=Cov+Cox;

CL(ON)=CL;

R(OFF)=large;

Cgd(OFF)=Cov;

CL(OFF)=CL+CoxCgb/(Cox+Cgb);

-   -   ˜CL+Cgb;

Cox>>Cgb, Csb.

For example, when the gated diode is OFF, the gate to body capacitanceis equal to Cox in series with Cgb. Adding this to CL gives theequivalent OFF load capacitance CL(OFF).

FIG. 1D shows an example of a side view cross-section of a first n-typegated diode 600 formed in Silicon-On-Insulator (SOI). The first n-typegated diode 600 comprises a gate insulator 620 formed between a gate 615(e.g., N+ doped polysilicon) and a p-well 630, a source diffusion region610, two STI regions 605 and 625, and an insulator 635. The p-well 630is formed above well boundary 636. The dopant concentration in p-well630 substantially controls the threshold voltage of the gated diode 600.

FIG. 1E shows an exemplary equivalent circuit used for modeling thefirst n-type gated diode shown in FIG. 1D. In FIG. 1E (with appropriatereference to FIG. 1D), R, Cov, Csb, Cox, Cgb, and CL are defined asabove, where the body of the gated diode 600 is the volume of the p-well630 under the gate 615 and between the source 610 and the STI region625. Additionally, Cbp is the capacitance between the body and theinsulator 635.

Exemplary equations suitable for modeling and that use therepresentative circuit of FIG. 1E are as follows, where “ON” indicatesthat the gated diode is turned on and “OFF” indicates that the gateddiode is turned off:

R(ON)=small;

Cgd(ON)=Cov+Cox;

CL(ON)=CL;

R(OFF)=large;

Cgd(OFF)=Cov+CsbCgb/(Csb+Cgb);

CL(OFF)=CL;

Cox>>Cgb, Csb>>Cbp.

FIG. 2A shows exemplary symbols used for a second n-type gated diode.Symbol 190 is an exemplary symbol for a second n-type gated diode shownin FIG. 2B. The same symbol 190 is used for both FIGS. 1A and 2A. FIG.2B shows an example of a side view cross-section of the second n-typegated diode 200 formed in a semiconductor. The second n-type gated diode200 comprises a gate insulator 220 formed between a gate 215 (e.g., N+doped polysilicon) and a p-well 230, a source diffusion region 210, twoSTI regions 205 and 225, an optional n isolation band 240, a p-substrate235, a “drain” diffusion region 245, and a coupling 250 thatelectrically couples source diffusion region 210 and drain diffusionregion 245. As described below, the dopant concentration in p-well 230substantially controls the threshold voltage of the gated diode 100.

FIG. 3 is a graph illustrating typical capacitance of the gatecapacitance Cgs (obtained by the derivative of charge with respect tovoltage, dq/dv) versus voltage between the gate and source (Vgs) for ann-type gated diode in bulk silicon, for a number of different gateareas. The gate capacitance Cgs includes the gate Metal OxideSemiconductor (MOS) capacitance formed by the dielectric under the gateand the overlap capacitance, Cov, between the gate and the source,excluding the gate to body and the source to body capacitances. Eachcurve corresponds to a gated diode having a certain gate area.

Curve ABCD shows the gate capacitance versus Vgs curve of a gated diodewith a threshold voltage, Vt, of 0.16V (point C) and certain gate area.When Vgs is above Vt, there is a substantial amount of charge stored inthe inversion layer (point A); and when Vgs is below Vt, the amount ofinversion charge is orders of magnitude smaller (point D). The gate tosource ON capacitance reaches maximum at point A. The maximum isapproximately 150 millivolts (mV) above Vt with Cg_gd(ON) of about 2.1femtofarads (fF). When Vgs is below Vt, the gate to source OFFcapacitance reaches minimum at point D, which is about 150 mV below Vtwith Cg_gd(OFF) of about 0.2 fF. Below Vt, the inversion charge andcapacitance are absent, and only the overlap capacitance between gateand source is present. The capacitance changes drastically around Vt(point C in FIG. 3) and its value levels off quickly to about 150 mVbeyond Vt. The threshold voltage (Vt_gd) can be controlled by the amountof implanted dopant, which is a key parameter in circuit design. Thoseskilled in the art should know that the amount of implant dopant cantrade off margins of signal, noise, and Vt variation. A low dopant levelgiving Vt_gd 50 to 100 mV can be beneficial to provide a good amount ofcharge and voltage for one-data (e.g., data that represents a “one”),and sufficient separation from ground noise.

In this disclosure, if it is not mentioned explicitly, a gated diode isassumed to be an n-type, as well as the associated NFET and PFET(transistors) in the corresponding circuits. For p-type gated diode, aswell as the associated PFET and NFET (transistors) in the correspondingcircuits, voltages and operations are complementary to the n-type case,and can be readily designed correspondingly, by someone who is skilledin the art.

Turning now to FIG. 4, a voltage boosting circuit 400 is shown using acapacitor 430 as a charge storage device. Amplifier 400 is coupled to asignal node 410, and has a capacitor 430 whose first terminal is coupledto the signal node 410 and whose second terminal is coupled to a setline 420. The signal node 410 has a capacitance 440 of CL, which is thelumped capacitance from the signal node 410, plus the couplingcapacitance and the total capacitance of the connecting circuits (ifthere is any capacitance) to the signal node. The capacitive load (CL)is not considered part of the amplifier 400.

During signal boosting, the set line voltage (Vs) on the set line 420 israised or boosted. Following the set line voltage, the source voltage ofthe signal node 410 is also therefore boosted by certain amount (denotedby VB), typically 50 percent to 100 percent of the supply voltage (VDD).The magnitude of the set line voltage (Vs) can be a predeterminedvoltage of a digital signal or its magnitude can be varied to give theamount of voltage boosting needed, as the boosted voltage on the signalnode 410 (after Vs is raised) depends on the magnitude of the set linevoltage, as well as the characteristic of the gated diode and the loadcapacitance CL.

Referring now to FIG. 4B, graphs are shown illustrating gain for thevoltage boosting circuit 400. The first graph, Vs, shows how the voltagevaries on the set line 420. The second graph shows how the voltage atthe point 401 would vary. As seen in FIG. 4B, the gain is about one forthe voltage boosting circuit 400. If the signal node 410 has a highvoltage, the output will be VB plus the high voltage (data one). If thesignal node 410 has a low voltage, the output will be VB plus the lowvoltage (data zero). The difference, dVin, is whatever difference existsbetween the data one and data zero voltages. Thus, the gain, which isdVout divided by dVin (the data one voltage minus the data zero voltage)is about one. Additionally, the flip point voltage, which is the voltageat which a decision is made as to whether a data one or data zero isseen, is relatively small if this circuit is used for signal detection.In other words, the signal margin of this circuit 400 is relativelysmall, and its current driving capability measured by dVout for drivingan output buffer or an inverter or a latch is relatively small comparedto a gated diode amplifier which will be described next.

FIGS. 5A, 5B and 5C illustrate exemplary principles of operation of agated diode amplifier 500. In FIG. 5A, a gated diode amplifier 500 isshown. Gated diode amplifier 500 is coupled to a signal node 510, andhas a gated diode 530 whose gate terminal (and therefore gate) iscoupled to the signal node 510 and whose source terminal (and thereforesource diffusion region) is coupled to a set line 520. The loadcapacitance CL(ON) and CL(OFF) for ON or OFF, respectively, includes thelumped total of external load capacitance CL and the internal equivalentcapacitance from the gated diode, as summarized in FIGS. 1C and 1E. Thecapacitive load (CL) is not considered part of the gated diode amplifier500. A control voltage Vs is connected to the source of the gated diode.To operate the gated diode amplifier 500, the voltage at the source ofthe gated diode is raised.

Let the gated diode gate to source ON capacitance be Cg_gd(ON), and OFFcapacitance be Cg_gd(OFF). Let Rc=Cg_gd(ON)/CL(ON) andrc=Cg_gd(OFF)/CL(OFF). The total load CL(ON) and CL(OFF) are lumpedcapacitance at the signal node to ground, and these may include the gateto source capacitance of a next stage FET, any stray capacitance on thesignal node to ground, as well as some internal device capacitance ofthe gated diode, e.g., the gate to body capacitance of the gated diodewhen it is OFF.

As shown in FIG. 5C, VL_HIGH is the voltage level for a “one” (e.g.,data one), and VL_HIGH>Vt_gd. There is a substantial amount of chargestored, given by (VL_HIGH−Vt_gd) Cg_gd(ON), represented by point a or Ain FIG. 3. The stored charge effectively causes a large capacitance, sothat the gated diode 530 may be represented as large capacitor 560. WhenVs is raised by VB, there is a large voltage increase at the gate of thegated diode. The maximum voltage is given by VL_HIGH+VB Rc/(1+Rc). Thefinal value Vg_f depends on the amount of charge stored and transferred,and Rc.

As shown in FIG. 5B, VL_LOW is the voltage level for a “zero” (e.g.,data zero), and VL_LOW<Vt_gd. When Vs is raised, the voltage at the gatestays almost zero since Cg_gd(OFF) (point D in FIG. 3) is much smallerthan CL(OFF). The charge stored effectively causes a small capacitance,so that the gated diode 530 may be represented as small capacitor 550.The voltage is given by VL_LOW+VB rc/(1+rc).

If Rc>>1, there is enough charge in the gated diode to transfer to theload CL(ON), without affecting the gated diode ON capacitancesignificantly (e.g. from point a to A to B in FIG. 3). The gated diodeis operating under the constrained charge transfer mode. For instance,VL_HIGH=0.6V (point a in FIG. 3), VB=1.2V, if final gate voltage(Vg_f)=1.4V, then the final Vgs of the gated diode=0.2V (point B in FIG.3). The charge transfer out from the gated diode to the load CL(ON) isthe area under the curve a-B in FIG. 3, represented by the area enclosedby a-B-B′-a′.

When Cg_gd(ON)>>CL(ON) and CL(OFF)>>Cg_gd(OFF), the output voltage atthe gate can be approximately by:Vout(1)=VL_HIGH+VB Rc/(1+Rc); andVout(0)=VL_LOW+VB rc/(1+rc).

As shown in FIGS. 4B and 6, let dVin be the difference of the gatevoltage between data zero and data one before Vs is raised, and dVout besuch difference after Vs is raised. Typically, VL_LOW=0. In the case ofa linear capacitor amplifier (see FIG. 4), the capacitance is constantthroughout for data zero and data one, dVin=dVout, so the gain is one.In the gated-diode case (see FIG. 6),

dVout = VL_HIGH + VB Rc/(1 + Rc) − (VBrc/(1 + rc) + VL_LOW);dVin = VL_HIGH − VL_LOW;gain = dVout/dVin; = 1 + (VB/VL_HIGH)[Rc/(1 + Rc) − rc/(1 + rc)] > 1.For example, VB=1.2V, VL_HIGH=0.6V, VL_LOW=0, rc=0.1, Rc=9, dVout=1.57V,dVin=0.6V, gain=2.62.

In addition to the gain advantage of a gated diode amplifier 500 ascompared to a linear capacitor voltage boosting circuit 400, the gateddiode amplifier 500 provides more margins for voltage distinction at theoutput stage. Without loss of generality, assume an inverter with itsinput connected to the signal node is used to detect data zero or dataone. The data zero and data one voltage levels of the linear capacitorcase and the gated diode case are shown in FIGS. 4 and 6, respectively.The zero to one flip point voltage would be selected mid-way between thevoltage outputs for the data zero and data one. The flip point voltageis the mid-point of dVout shown in FIGS. 4 and 6 for the cases of thelinear capacitor and the gated diode case, respectively. The detectionmargin can be defined as dVout/2. Since the gated diode amplifier 500has a much higher dVout, it has a much higher margin of error toseparate data zero and data one, as compared to a linear capacitorvoltage boosting circuit 400. Further, the output current of theinverter are determined by its input overdrive voltage, which is|Vout−Vt| for both P-type FETs (PFETs) and N-type FETs (NFETs). Suchoverdrive equates to dVout for both data zero and data one. Since dVoutis much larger for the gated diode case, output current of the gateddiode amplifier 500 is higher and its speed is faster compared to thelinear capacitor voltage boosting circuit 400.

Let Vg_f be the final gate voltage. The final voltage across the gateddiode is (Vg_f−VB) and it is less than the initial voltage VL_HIGH. LetVxfer be the decrease of voltage across the gated diode,Vxfer=VL_HIGH+VB−Vg_f. Part or all of the charge in the gated diode istransferred to the load as the load voltage increases from VL_HIGH toVg_f. The charge transfer to the load CL(ON) Qxfer is given byQxfer=(Vg _(—) f−VL_HIGH)CL(ON)=(VB−Vxfer)CL(ON).

Qxfer is given by the area under the capacitance-Vgs curve betweenVL_HIGH and (Vg_f−VB) (points a-B in FIG. 5). Vg_f can then bedetermined graphically or numerically from Qxfer. For example, initiallyVgs(at point a)=VL_HIGH, charge transfer is represented by moving frompoint a to B, and the final voltage is given by point B. Vg_f=VB+Vgs(atpoint B). The gain can then be calculated bygain=dVout/dVin=(Vg _(—) f−VB rc/(1+rc))/VL_HIGH.

For completeness, for small Rc, all the charge stored in the gated diodeis transferred to the load, the complete charge transfer mode,represented by curves B-D, A-D or a-D in FIG. 3. The gated diode isturned OFF. The final voltage at the gate Vg_f is given by thefollowing:

Qxfer = (VL_HIGH − Vt_gd)Cg_gd(ON) = (Vg_f − VL_HIGH)CL(ON);Vg_f = (VL_HIGH − Vt_gd)Cg_gd(ON)/CL(ON) + VL_HIGH;Vg_f(VL_HIGH − Vt_gd)Rc + VL_HIGH;andgain = dVout/dVin = Vg_f − VBrc/(1 + rc))/VL_HIGH = 1 + Rc(1 − Vt_gd/VL_HIGH) − VB/VL_HIGHrc/(1 + rc) ∼ 1 + Rc(1 − Vt_gd/VL_HIGH)(when  rc ⪡ 1).

When the source voltage returns to ground, the charge that transferredout of the gated diode to CL(ON) will return back to the gated diode,the gate voltage will return to the pre-boosted value, and the totalcharge stored in the gated diode and its load is conserved before andafter the read.

Referring now to FIG. 7, a top view is shown of two of the first n-typegated diodes formed in a semiconductor. Two planar gated diodes areformed by source area 711, 712 of a diffusion 710 underneath apolysilicon gate area 730 that forms a gate 731. One gated diode isformed by source region 711 and gate 731, while another gated diode isformed by source region 712 and gate 731. Contacts 720, 721, 722 areprovided to contact the source areas 711, 712 and the gate 731respectively. The two gated diodes can be connected in parallel,implementing a structure as shown in FIG. 2B, by external metalconnections and contacts. If only one side of the source area 711 or 712is present, then a single gated diode is formed with gate 731 and source711, or with gate 731 and source 712, implementing structures as shownin FIG. 1B and FIG. 1D.

The variables L, W, and Lov are the following: L is the gate length, Wis the gate width, and Lov is the overlap length between the gate andthe source. The gain is as follows:Gain˜1+[Cox/(Cox+CL)−Cov/(Cov+CLoff)] VB/VL_HIGH.VB, VL_HIGH, Cox, Cov have been defined earlier. To improve gain, Coxshould be large and Cov should be small. It is recommended that L>>Lminand W>>Wmin, where Lmin and Wmin are the minimum sizes determined by thetechnology being used. A reasonably sized Cox is approximately two toten times CL. Selection of L>>Lmin, W>>Wmin and a reasonably sized Coxhas exemplary benefits of the following:

reduced dL induced Vt fluctuation;

reduced short channel effect;

the carrier transit time<a Resistance-Capacitance (RC) requirement; and

increased Cox/Cov=L/Lov, hence achieving higher GAIN.

The following are also recommended: (1) low Vt=50–100 mV, which storesmore charge, allowing small signal (e.g., 200 mV) to be sensed; and (2)low dose ion implantation and low background doping concentration, whichcause less dN/N dopant fluctuation, less change in oxide thickness(dTox) fluctuation effect on Vt, and smaller Vt fluctuation.

The precision of the Vt of the gated diode helps allow a gated diodeamplifier to detect small signals accurately. It is desirable to havethe Vt midway between the low and high of the small signal voltage todistinguish between the data zero and data one. More importantly, the Vtvariation of the gated diode should be small over various manufacturingprocess and wafer variations, in order to avoid giving false zeroindications when Vt shifts up, and false one indications when Vt shiftsdown. The allowable percentage variation in Vt of the gated diode shouldbe even smaller than that allowed for logic gates due to the smallmagnitude of the signal. The gated diode can be designed with low Vt andlow dopant concentration, which would provide minimal Vt variation andmaximal signal detection separation. The short channel or roll-offeffect for short channel logic devices is typically not an issue becausethe gated diode has no drain voltage to induce Drain-Induced BarrierLowering (DIBL) effect.

Since the gated diode should be a certain size to achieve the requiredgated diode capacitance to load capacitance ratio to achieve therequired gain as described above, the channel length of gated diode isnot necessarily of minimal channel length like the rest of the FETdevices used for logic. In order to maximize the gain, it is beneficialif the ratio of the gated diode ON and OFF capacitance is as highestpossible. Consequently, the capacitance Cox to the gate to sourceoverlap parasitic capacitance Cov should be as large as possible for agiven gate capacitance Cox. Since Cox/Cov=L/Lov, L should be made aslarge as possible provided that the threshold profile variation and theRC delay for carrier transport is within certain requirements known tothose skilled in the art.

Sense Amplifier Circuits Using Gated Diodes

Turning now to FIG. 8, a sense amplifier circuit 875 is shown coupled toa signal line 810 adapted to carry a small signal, Vi. The senseamplifier circuit 875 comprises control circuitry 890, a gated diodeamplifier 800, and an optional output device 860. The gated diodeamplifier 800 comprises an isolation device 845 and a gated diode 830.The optional output device 860 is typically a latch or buffer. In theexample of FIG. 8 and in the following examples, the output device 860is an inverter, which is one type of buffer. However, other types oflatches or buffers may be used in place of the inverter, for those whoare skilled in the art. The output device 860, an exemplary invertercomprises a PFET 865 and an NFET 870. The sense amplifier circuit 875 iscoupled to an output line 880 and produces an output signal, Vout. Thecontrol circuitry 890 is coupled, in the illustration of FIG. 8, to acontrol terminal of the isolation device 845 through an isolation devicecontrol line 881 and to the set line 820. The input capacitance, Cin, isrepresented by capacitor 850. The sense amplifier circuit 875 is coupledto a signal line 810 having a small signal, Vi, placed thereon. Thesignal line has a capacitance, Cin, illustrated by capacitor 850 anddescribed in more detail below.

The isolation device 845 and other PFET and NFET devices describedherein can be considered to be switches, where a voltage that is withina predetermined voltage range and that is applied (e.g., through acontrol line) to control terminals of the devices causes an electricalconnection between a first terminal of the devices and between a secondterminal of the devices. In the example of the isolation device 845, theappropriate voltage on the control terminal will electrically couple asmall signal, Vi, to a first terminal of the gated diode 830 and tocontrol terminals on the PFET 865 and the NFET 870. The switch (e.g.,the isolation device 845) is turned ON (e.g., enabled). Similarly, avoltage that is not within the predetermined voltage range and that isapplied (e.g., through the isolation device control line 881) to thecontrol terminal of the isolation device 845 causes an electricaldisconnection between the first terminal of the isolation device 845 andthe second terminal of the isolation device 845. The switch (e.g., theisolation device 845) is turned OFF (e.g., disabled). The thresholdvoltage of the isolation device 845, ground voltage and the power supplyvoltage (VDD) typically define the predetermined voltage range forP-type and N-type switches, such as PFETs and NFETS. The PFET 865 andNFET 870 may also be considered to be switches.

In FIG. 8, an isolation device 845 is added to the basic form of gateddiode amplifier 500 (see FIG. 5A) between the gated diode 830 and thesignal line 810 to handle signal lines with high capacitive load, asindicated by Cin and capacitor 850. The gate of the gated diode 830 isisolated from the signal line 810 and signal source (not shown) so thatthe signal line 810 and the signal source do not load the sensing node801. The gated diode 830 is used to create a non-linear boosting of thevoltage at the sensing node 801, as described earlier. The isolationdevice 845 isolates the signal line from the sensing node, where theload CL is only for the node Vgd (e.g., the sensing node 801) includingthe input capacitance of the output device 860 and the parasiticcapacitances of the gated diode 830 and the isolation device 845.

The isolation device 845 can be implemented either with a carefullychosen constant voltage Vc applied at the gate, such as described in“Amplifiers Using Gated Diodes,” already incorporated by referenceabove, or with a separate digital signal to switch the isolation device845 to ON to pass the small signal, Vi, to the gated diode, and OFF toisolate the gated diode from the signal source, typically after thesmall signal Vi is sampled and stored in the sensing node 801 during theON phase. The present disclosure describes using a digital signal tocontrol the voltage on the control terminal of the isolation device 845.A digital signal is a signal controlled to turn the isolation device8450N and to turn the isolation device 845 OFF, typically with twodifferent voltages.

By controlling the voltage on the control terminal of isolation device,sensing speed may be increased as compared to using a constant voltagefor the isolation device 845. Additionally, when there is a constantvoltage on the control terminal of the isolation device 845 and thevoltage at sensing node 801 is a predetermined voltage, the isolationdevice can be turned off. However, even when off, there may be a smallamount of leakage (e.g., voltage, current, or both) from the signal line810 to the point 801, e.g., when a FET is used as the isolation device845. When the control circuitry 890 is used to digitally switch (e.g.,using the isolation device control line 881) the voltage on the controlterminal of the isolation device 845, this small amount of leakage canbe reduced or eliminated.

In the example of FIG. 8, the control circuitry 890 raises voltage of asampling signal, bVs, during a sampling operation, and produces asampling signal, bVs, on the control terminal, connected to the controlline 881. The sampling signal, bVs, is the complement of the set signal,Vs. After the sampling operation, the control circuitry 890 will lowerthe sampling signal, bVs. There are a number of operations that a senseamplifier circuit, such as sense amplifier circuit 875, typicallysupports. In a sampling operation, the small signal (Vi) is sampled sothat a voltage corresponding to the small signal is stored and thentypically isolated at sensing node 801 after the gated diode 845 isturned OFF. In a sensing operation, the set signal, Vs, on the set line820 is raised (e.g., by the control circuitry 890) so that anappropriate sensed output corresponding to the sampled voltage, isolatedat sensing node 801, is output as output signal, Vout. As described inmore detail below, these two operations, namely sampling and set, may beseparated by a certain time period or may overlap.

In general, as described earlier, the magnitude of Vs can be apredetermined voltage of a digital signal or it can be made adjustableto give the desired voltage boosting needed for the gated diodeamplifier.

Referring now to FIG. 9, a number of waveforms are shown for the senseamplifier circuit of FIG. 8. FIG. 9 also compares waveforms of a gateddiode sense amplifier 875 with waveforms produced by a typical senseamplifier. A waveform for the small signal, Vi, on the signal line 810is shown first. A waveform for the set signal, Vs, is shown. Note thatthe sampling signal, bVs, is the complement of Vs. A waveform forsensing node 801, which is the gated diode output, is shown next. Awaveform for the output signal, Vout, on output line 880 is shown. Forcomparison purposes, the final waveform shows an output signal, Vout,from conventional sense amplifier circuit.

It can be seen in FIG. 9 that the sense amplifier circuit 875 with agated diode amplifier 800 produces an output signal, Vout, faster than aconventional sense amplifier circuit.

FIG. 10 is an example of another sense amplifier circuit 1075 using agated diode amplifier and control circuitry adapted to control certainelements of the amplifier circuit. Sense amplifier circuit 1075 is showncoupled to a signal line 1010 adapted to carry a small signal, Vi. Thesense amplifier circuit 1075 comprises control circuitry 1090, a gateddiode amplifier 1000, and an optional output device 1060. The gateddiode amplifier 1000 comprises an isolation device 1045 and a gateddiode 1030. The optional output device 1060 comprises an inverter havinga PFET 1065 and an NFET 1070. The sense amplifier circuit 1075 iscoupled to an output line 1080 and produces an output signal, Vout. Thecontrol circuitry 1090 is coupled to a control terminal of the isolationdevice 1045 through an isolation device control line 1081 and to the setline 1020. The input capacitance, Cin, is represented by capacitor 1050.In this example, the isolation device control line 1081 has a samplingsignal thereon, SMPL, and the set line has a set signal, SET, thereon.

For an operation of sampling, one exemplary technique used is asfollows. The control circuitry 1090 adjusts the following signals, where“high” is a predetermined high voltage and a “low” is a predeterminedlow voltage:

SMPL=high; and

SET=low,

such that the isolation device 1045 is ON. The small signal, Vi, issampled and stored at node Vgd at point 1001. The small signal is storedat point 1001 even after SMPL goes to zero.

For an operation of sensing, one exemplary technique used is as follows.The control circuitry 1090 adjusts the following signals:

SMPL=low;

SET goes from low to high to boost Vgd.

If Vgd is low, Vout stays high. On the other hand, if Vgd is high, Voutfalls to low. As described above in reference to FIGS. 8 and 9, thesignals of SMPL and SET can be complementary. Alternatively, the SMPLsignal can be turned off sooner than the SET signal goes high.

As another example, assume that the sampling signal SMPL is thecomplement of SET (bSET) logically, such that SET is a delayedcomplement of SMPL. During a sampling operation, when SMPL=high (orbSET=high) and SET=low, the small signal is sampled, so that the smallsignal passes through the isolation device 1045 and is held temporarilyat Vgd (the sensing node 1001) by the capacitance of the gate of thegated diode 1030 and the gates of the PFET 1065 and NFET 1070 of theoutput device 1060. At the end of the sampling operation, SMPL=low andthe sampled small signal voltage is held by the capacitance at thesensing node 1001 as Vgd as the isolation device 1045 is turned OFF.During the sensing operation, the set signal, SET, switches to high(SMPL=bSET=low), and the isolation device 1045 has been turned OFF. Forcorresponding data one, the gated diode 1030 then boosts the temporarilyheld small signal Vgd to full logic swing, and for data zero, Vgdremains low. The full logic swing of Vgd corresponding to data zero andto data one would then be output via the output device 1070, and avoltage corresponding to the output signal 1080 can then be stored orpassed to subsequent logic stage.

In general, as described earlier, the magnitude of SET signal can be apredetermined voltage of a digital signal or it can be made adjustableto give the desired voltage boosting needed for the gated diodeamplifier.

After the sensing operation, the SET signal on set line 1020 returns tolow and the SMPL signal on the isolation device control line 1081returns to high to complete a SMPL and SET sensing cycle. Even underheavy input line loading, the sense amplifier circuit 1075 can achievevery fast switching time, since the gated diode 1030 loading can belimited to a very small amount equivalent to about the loading ofminimum feature size. For use in a bank or a large number of senseamplifiers, the two SMPL and SET signals can be shared among many gateddiode amplifiers 1000.

The exemplary sense amplifier circuit 1075 works in a way that theisolation device 1045 can be controlled digitally, without the need toset Vc precisely. As a result, the sense amplifier circuit 1075 isrobust and tolerant to Vt variation, voltage, temperature and processvariation. The isolation device 1045 can be used as part of amultiplexer (MUX) network to control the signal flow of the small signalinto the gated diode amplifier from one of many sources.

Turning now to FIG. 11, another example of a sense amplifier circuit1175 is shown. Sense amplifier circuit 1175 is shown coupled to a signalline 1110 adapted to carry a small signal, Vi, and having an inputcapacitance of Cin, as illustrated by capacitor 1150. The senseamplifier circuit 1175 is also shown coupled to an optional keeper 1140,which comprises two inverters 1130 and 1140. The sense amplifier circuit1175 comprises control circuitry 1190, a gated diode amplifier 1100, andan optional output device 1160. The gated diode amplifier 1100 comprisesan isolation device 1145 and a gated diode 1130. The optional outputdevice 1160 comprises an inverter having a PFET 1165 and an NFET 1170.In this example, a control terminal of the PFET 1165 is coupled to thecontrol circuitry 1190 through a PFET control line 1182. The controlcircuitry 1190 creates a PFET control signal, bPC, on the PFET controlline 1182. The sense amplifier circuit 1175 is coupled to an output line1180 and produces an output signal, Vout, on an output line 1180. Thecontrol circuitry 1190 is coupled to a control terminal of the isolationdevice 1145 through an isolation device control line 1181 and is coupledto the set line 1120. In this example, the control circuitry 1190creates a sampling signal, SMPL, on the control terminal 1181 and a setsignal, SET, on the set line 1120. There is a sensing node 1101 in FIG.11.

The keeper 1140, made up of two small inverters 1130 and 1140 formingone example of a latch, is optional. The keeper 1140 is used forholding, if needed, the output voltage of the dynamic node, Vout, on theoutput line 1180 for an indefinite period of time until the output nodeVout is precharged again.

An exemplary technique for operating the sense amplifier circuit 1175 isas follows. For a sampling operation, the following signal voltages arecreated by control circuitry 1190:

SET=bPC=low;

SMPL=high; and

Vout is precharged to high.

The SMPL signal causes the isolation device 1145 to be ON. The smallsignal, Vi, on signal line 1110 is sampled and stored at the sensingnode 1101, Vgd. The sampled small signal is stored even after SMPL goeslow, turning OFF the isolation device 1145.

During a sensing operation, the following signal voltages are created bythe control circuitry 1190:

SMPL=low;

bPC=high; and

SET goes from low to high to boost Vgd.

If the voltage at the sensing node 1101, Vgd, is low, then the outputsignal, Vout, stays high. Conversely, if Vgd is high, the NFET 1170turns ON and discharges the precharged high voltage on the output line1180 to low, so Vout falls to low. The signals SMPL and SET can becomplementary or SMPL can be turned off sooner than SET goes high. Thesignals bPC and SET can be same signal, if desired.

In general, as described earlier, the magnitude of SET signal can be apredetermined voltage of a digital signal or it can be made adjustableto give the desired voltage boosting needed for the gated diodeamplifier.

Turning now to FIG. 12, a sense amplifier circuit 1275 is shown. Senseamplifier circuit 1275 is shown coupled to N signal lines 1210-1 through1210-N, each of which is adapted to carry a small signal, Vi, and has aninput capacitance of Cin, as illustrated by capacitors 1250-1 through1250-N. In general, the N capacitors 1250-1 through 1250-N can be ofdifferent capacitance. The sense amplifier circuit 1275 is also showncoupled to an optional keeper 1140, which operates as described inreference to FIG. 11. The sense amplifier circuit 1275 comprises controlcircuitry 1290, a gated diode amplifier 1200, and an optional outputdevice 1260. The gated diode amplifier 1200 comprises N isolationdevices 1245-1 through 1245-N and a gated diode 1230. The optionaloutput device 1260 comprises an inverter having a PFET 1265 and an NFET1270. In this example, a control terminal of the PFET 1265 is coupled tothe control circuitry 1290 through a PFET control line 1282. The controlcircuitry 1290 creates a PFET control signal, bPC, on the PFET controlline 1282. The sense amplifier circuit 1275 is coupled to an output line1280 and produces an output signal, Vout, on an output line 1280. Thecontrol circuitry 1290 is coupled to control terminals of the isolationdevices 1245 through isolation device control lines 1281-1 through1281-N and is coupled to the set line 1220. In this example, the controlcircuitry 1290 creates sampling signals, SMPL-1 through SAMP-N, on thecontrol terminals 1281 and a set signal, SET, on the set line 1220.There is a sensing node 1201 in FIG. 12.

An exemplary technique for operating the sense amplifier circuit 1275 isas follows. For a sampling operation, the following signal voltages arecreated by control circuitry 1290:

SET=bPC=low;

SMPLi=high (e.g., a selected one of the N isolation device control lines1281 is high, the rest are low); and

Vout is precharged to high.

The high voltage on one of the selected isolation device control line1281 causes a corresponding one of the isolation devices 1245 to be ON.The small signal, Vi, on a corresponding signal line 1210 is sampled andstored at the sensing node 1201, as Vgd. The stored small signal isstored even after SMPLi goes low.

During a sensing operation, the control circuitry 1290 creates thefollowing signal voltages:

SMPLi=low, for i=1, . . . , N (all SMPL=low);

bPC=high; and

SET goes from low to high to boost Vgd.

If the voltage at the sensing node 1201, Vgd, is low, then the outputsignal, Vout, stays high. Conversely, if Vgd is high, the NFET 1270turns ON and discharges the precharged high voltage on the output line1280 to low, so Vout falls to low. The signals SMPLi and SET can becomplementary or SMPLi can be turned off sooner than SET goes high. Thesignals bPC and SET can be same signal, if desired.

In general, as described earlier, the magnitude of SET signal can be apredetermined voltage of a digital signal or it can be made adjustableto give the desired voltage boosting needed for the gated diodeamplifier.

Turning now to FIG. 13, a sense amplifier circuit 1375 is shown. Senseamplifier circuit 1375 is shown coupled to a signal line 1310, which isadapted to carry a small signal, Vi, and has an input capacitance ofCin, as illustrated by capacitor 1350. The sense amplifier circuit 1375is also shown coupled to an optional keeper 1140, which operates asdescribed in reference to FIG. 11. The sense amplifier circuit 1375comprises control circuitry 1390, a gated diode amplifier 1300, and anoptional output device 1360. The gated diode amplifier 1300 comprises anisolation device 1345 and a gated diode 1330. The optional output device1360 comprises an inverter having a PFET 1365 and an NFET 1370. In thisexample, a control terminal of the PFET 1365 is coupled to the controlcircuitry 1390 through a PFET control line 1382. The control circuitry1390 comprises isolation device control line 1383. The control circuitry1390 creates a PFET control signal, bPC, on the PFET control line 1382.The sense amplifier circuit 1375 is coupled to an output line 1380 andproduces an output signal, Vout, on an output line 1380. The controlcircuitry 1390 is coupled to the set line 1320. In this example, thecontrol circuitry 1390 creates a set signal, SET, on the set line 1320.There is a sensing node 1301 in FIG. 13.

An exemplary technique for operating the sense amplifier circuit 1375 isas follows. For a sampling operation, the following signal voltages arecreated by control circuitry 1390:

SET=bPC=low; and

Vout is precharged to high.

In FIG. 13, the sense amplifier circuit 1375 precharges Vout high byusing the control signal bPC. During the sampling operation, the highoutput of the output signal, Vout, is fed back to the control terminalof the isolation device 1345 through the isolation device control line1383 so the isolation device 1345 is turned ON, as a result the smallsignal Vi appears at the sampling node 1301 (e.g., signal Vgd). Afterpre-charging (bPC=high), the control signal SET is set to high and thegated diode 1330 boosts the voltage at the sampling node 1301corresponding to data one or data zero, as described above in referenceto the gated diode amplifier operation. The boosted voltage in turnwould switch the pre-charged output device 1360 accordingly during asensing operation.

Thus, during the sensing operation, the control circuitry 1390 createsthe following voltage signals:

bPC=high; and

SET goes from low to high to boost Vgd.

If Vgd is low, Vout stays high, as the isolation device 1345 remains ONbecause Vi and Vgd are low. If Vgd is high, Vout falls to low and theisolation device 1345 turns off, as Vout is routed through isolationdevice control line 1383 to the control terminal of the isolation device1345. The voltage at node Vout is held constant by using the keeper 1140until the next precharge operation when SET and bPC go low again. Thesignals bPC and SET can be the same signal.

In general, as described earlier, the magnitude of SET signal can be apredetermined voltage of a digital signal or it can be made adjustableto give the desired voltage boosting needed for the gated diodeamplifier.

Therefore, during the sensing operation, SET=high after being raised. Ifthe small signal corresponds to a data zero, the sampling node 1301,having signal Vgd, remains low and output signal remains high. On theother hand, if the small signal corresponds to a data one, Vgd isboosted high, and the output device 1360 switches the output signal tolow and also turns OFF the isolation device (id). The next samplingoperation will start a new cycle. In this circuit arrangement, the twocontrol signals bPC and SET can be the same signal, reducing to only asingle control signal. The operation is divided into a samplingoperation (e.g., bPC=SET=0) and a sensing operation (e.g., SET=1). Anoptional keeper circuit, such as keeper 1140, may be needed to hold theoutput voltage of the output line 1380 due to leakage current throughthe NFET 1370 of the output device 1360.

The feedback connection 1383, from the output node Vout to the controlterminal of the isolation device, forms a unique “self-sampling” circuittopology comprising the gated diode amplifier 1300 and an prechargedinverter 1360 with precharge signal 1382 such that only one controlsignal is needed, namely the set signal SET which can be used for thePFET precharge control signal bPC.

Referring now to FIG. 14, a memory 1400 is shown. Memory 1400 comprisesa memory array 1410, a number of wordline drivers 1480, and a sensingblock 1405. The memory array 1410 comprises a number of bitlines 1415,and a number of wordlines 1420. At the intersection of each bitline 1415and wordline 1420 is a memory cell 1425. One of the bitlines 1415 is atrigger bitline 1440 and one of the wordlines 1420 is trigger wordline1430. At the intersection of the trigger bitline 1440 and the triggerwordline 1430 is a trigger memory cell 1435. The sensing block 1405comprises an array 1495 of gated diode sense amplifiers 1470, andcontrol circuitry 1490. Control circuitry 1490 comprises two inverters1451 and 1452 to generate the complementary control signals SET and SMPLand a trigger circuit 1465. The control circuitry 1490 is coupled to aset line 1445 and produces a SET signal on the set line 1445. Similarly,the control circuitry 1490 is coupled to an isolation device controlline 1450 and produces a SMPL signal on the isolation device controlline 1450. The gated diode sense amplifiers 1470 are sense amplifierssuch as those previously described (e.g., based on the gated diodeamplifier 1000 of FIG. 10). It shall be understood that the use of thecontrol signal SMPL is optional, as in the case when the gated diodesense amplifier of FIG. 13 is used, the isolation device is internallycontrolled by the sense amplifier output, and so the control signal SMPLis not needed. The trigger circuitry 1465 is used to generate a triggersignal 1460, as described below.

To generate the SET signal, for example for the gated diode senseamplifier shown in FIG. 13; or to generate the SMPL and SET signals forexample for the gated diode sense amplifiers shown in FIGS. 10–12, thememory 1400 operates in an exemplary embodiment as follows. There is anextra bitline 1415 that is the trigger bitline 1440 and there is anextra wordline 1420 that is trigger wordline 1430. The trigger memorycell 1435 is at one end of the wordline 1430 from the correspondingwordline driver 1480 and has a known state (e.g., data one or data zero)stored. The extra trigger bitline 1440, wordline 1430, and memory cell1435 are built into the memory array 1410 to emulate hardware variationsdue to devices, voltages, processes, and temperature variations. Thesewordlines, bitlines and memory cells:

-   -   take into account worst-case signal delay in bitlines;    -   take into account worst-case signal delay in wordlines; and    -   take into account activation and read latency of a memory cell.

Exemplary trigger signaling is as follows: trigger wordline 1430 fromlow to high to activate the trigger memory cell 1435. If the triggermemory cell 1435 stored a data one, a known state, the trigger bitline1440, once precharged high, will go low. Exemplary trigger circuitry1465 can be an inverter or an inverter with adjustable delay. Thetrigger circuitry 1465 generates an output, Vout, that goes high andthat is used by the control circuitry 1490 to generate the SET signal onthe set signal line 1445 for the gated diode sense amplifiers 1470. Ifneeded, the control circuitry 1490 can also generate the SMPL signal onthe isolation device control line 1450 for the gated diode senseamplifiers 1470.

FIGS. 15A and 15B show two graphs of waveforms determined by simulatinga gated diode sense amplifier circuit 1075 of FIG. 10. FIG. 15A showssensing a high voltage for a data one, and FIG. 15B shows sensing a lowvoltage for a data zero. The voltage waveforms of the voltages for theSMPL signal, SET signal, small input signal, sensing node, and outputsignal versus time are shown. First, during a sampling operation, theSMPL signal is high to turn on the isolation device, so that the smallsignal is applied via the isolation device to the gate of the gateddiode (e.g., at the sensing node voltage of Vgd). At a certaindetermined time, the SMPL signal is brought low (e.g., by controlcircuitry) and the isolation device is therefore turned off. The sampledvoltage is stored in the gated diode as Vgd. At about the same time theSMPL signal goes low and the isolation device is turned OFF, the SETsignal is triggered to boost the sensing node voltage, Vgd.

In the case of data one (the graph shown in FIG. 15A), the sampledsensing node voltage is around 240 millivolts (mV) (e.g., 20 percent ofVDD), and the sensing node voltage is boosted (with the rise of the SETsignal) to a high voltage of 1.29V with a supply voltage (VDD) of 1.2V,with a fast rise time of 20 picoseconds (ps). The boosted high voltageat the gates of the PFET and NFET of the output device turns the PFETOFF and the NFET ON and causes the output signal to fall low with a fastfall time. In the case of data zero (the graph shown in FIG. 15B), thesensing node voltage (Vgd) stays low (around 90 mV) when the SET signalis triggered. The output signal of the output device stays high. Thevoltage gain achieved is (1.29−0.09)/(0.24−0)=5. The data zero or dataone output signal can then be stored or passed to subsequent logicstages.

After the sensing operation, the SET signal is driven low and the SMPLsignal is driven high to complete the sampling and sensing operations.The driving of the SET signal low and the SMPL signal high is not shownin FIG. 15A or 15B. Even under heavy input line loading, the senseamplifier circuit achieves very fast switching time, such as 20 ps asshown, since loading of the gated diode can be limited to a very smallload equivalent to about the loading of minimum feature size.

Turning now to FIGS. 16A and 16B, these figures show two graphs ofwaveforms determined by simulating a gated diode sense amplifier circuit1375 of FIG. 13. FIG. 16A shows sensing a high voltage for a data one,and FIG. 16B shows sensing a low voltage for a data zero. The voltagewaveforms of the voltages for the SET signal, small input signal,sensing node, and output signal versus time are shown. There is no SMPLcontrol signal in this case, and SET is used for bPC also. So, there isonly one control signal. First, during a sampling operation, the bPCsignal, which is the same as the SET control signal, is low to turn onthe PFET of the output device (see FIG. 13). The output signal isprecharged high and so the isolation device is ON as the output voltageis fed back to the control terminal of the isolation device. The smallsignal is applied via the isolation device to the gate of the gateddiode (e.g., at the sensing node voltage of Vgd). The sampled voltageappears in the gated diode as Vgd. At a certain determined time, the SETsignal is triggered to high to boost the sensing node voltage, Vgd, bythe gated diode.

In the case of data one (the graph shown in FIG. 16A), the sampledsensing node voltage is around 240 mV (e.g., 20 percent of VDD), and thesensing node voltage is boosted (by the rise in the SET signal) to ahigh voltage of about 1.1V with a supply voltage (VDD) of 1.2V, with afast rise time of 20 picoseconds (ps). The boosted high voltage (Vgd) atthe gates of the NFET of the output device turns the NFET ON and causesthe output signal to fall low with a fast fall time. The isolationdevice is then turned OFF as its control terminal voltage is low. In thecase of data zero (the graph shown in FIG. 16B), the sensing nodevoltage (Vgd) stays at almost zero volts after the SET signal istriggered high. The output signal of the output device stays high. Thevoltage gain achieved is (1.1−0.0)/(0.24−0)=4.6. The data zero or dataone output signal can then be stored or passed to subsequent logicstages.

After the sensing operation, the SET signal is driven low to completethe sampling and sensing operations. The driving of the SET signal lowis not shown in FIG. 16A or 16B.

Referring now to FIGS. 17A through 17C, these figures show a number ofwaveforms for a memory, such as memory 1700 of FIG. 17. In the examplesof FIGS. 17A through 17C, the memory has two wordlines, WLw and WLr(e.g., wordlines 1415 of FIG. 14), coupled to each memory cell (e.g.,memory cells 1425 of FIG. 14). WLw is a wordline used for writinginformation into a memory cell, while WLr is a wordline used to readinformation from a memory cell. Both of these wordlines are coupled tothe set line of a memory cell. FIG. 17B shows a Primary Sense Amplifier(PSA) using one of the gated diode sense amplifier circuits describedabove, and FIG. 17C shows a Secondary Sense Amplifier (SSA) using one ofthe sense amplifier circuits described above. In FIGS. 17A through 17C,a “1” means that a “data one” is being accessed and a “0” indicates thata data zero is being accessed.

In FIG. 17A, a memory cell is written by driving the WLw signal high,which stores (in the gated diode) a voltage of about 0.6 volts. Thememory cell is read by driving the WLr signal high. The “cell read for1” indicates what the internal voltage for the memory cell is during aread of a data one. Similarly, the “cell write for 0” indicates what theinternal voltage for the memory cell is during a write of a data zero.FIG. 17A also shows a memory cell write cycle for a data zero and amemory cell read cycle for a data zero.

FIG. 17B shows the SET signal for a gated diode sense amplifier circuitof a PSA. The PSA output (e.g., the output signal of the sense amplifiercircuit) goes high as the SET signal goes low. In this example, the PSAhas an input signal of the signal on the bitline. The PSA internalvoltage, which is the voltage at the sensing node of a gated diode senseamplifier, is shown for a read of a data one and a read of a data zero.The PSA has an output signal of a global bitline.

FIG. 17C shows a Secondary Sense Amplifier (SSA) that has an inputsignal of the voltage of the global bitline. The SSA internal voltage,which is the voltage at the sensing node of a gated diode senseamplifier, is shown for a read of a data one and a read of a data zero.The output of the SSA is also shown.

The gated diode sense amplifier circuits described above are designedand operate in a way that the isolation device can be controlleddigitally, without the need to set Vc precisely to certain valuedepending on Vt of the isolation device and the small signal magnitude.As a result, the gated diode sense amplifier circuits become more robustand tolerant to Vt variation, voltage, temperature and processvariation. The isolation device operates digitally to control the signalflow of the small signal into the gated diode amplifier.

Sense Amplifier Latch Circuits

In another aspect of the present invention, latch circuits with built-insense amplifiers are described. These latch circuits can achieve thecharacteristics of high performance, high data throughput, low powerdissipation, and resiliency to variations in the fabrication processsequence. These characteristics are becoming increasingly difficult toachieve as silicon CMOS technology advances. The sense amplifier latchcircuits described herein can be used to efficiently detect slow-movingoutputs of logic, memory, or long bus lines. In addition to improvedoperation, these latch circuits are much simpler to design than existingcircuits that accomplish similar functions.

By adding an appropriate precharge device and clock inputs to the senseamplifier circuits described above, the resulting circuit can operate asa latch. The clocked nature of the latch circuit allows for a pipelinedarchitecture within the sense amplifier circuit, which maximizes datathroughput. The sense amplifier latch circuits herein can be considered,for instance, to be dynamic latches that can provide signalamplification and improved data rates. In addition, the sense amplifierlatch circuits can be designed to be extremely insensitive to processvariations. The sense amplifier latch circuits are applicable to anyFET-based device technology that can provide transistor switches and atwo terminal semiconductor device having a non-linear capacitance.

FIG. 18 shows an example of a sense amplifier latch circuit 1875. Senseamplifier latch circuit 1875 comprises a precharge device 1820, a gateddiode sense amplifier 1800, and an output buffer 1860. The gated diodeamplifier 1800 comprises a pass-gate 1845 and a gated diode 1830. Theoutput buffer 1860 comprises a PFET 1865 and an NFET 1870. The senseamplifier latch circuit 1875 is coupled to an input 1810 (e.g., a signalline) and produces an output 1880. RC delay element 1815 is an exemplaryelement indicating that the input 1810 is a slow moving signal thatneeds amplification.

In FIG. 18, the sense amplifier latch circuit 1875 can be used to sensea signal on the input 1810 as the signal falls from the power supplyvoltage, VDD, toward a zero potential. The pass-gate 1845 is a switchthat is used to control dataflow. The gated diode 1830 is a device usedto perform signal amplification. A precharge switch, the prechargedevice 1820, is used to control the input signal on input 1810. The PFET1865 and NFET 1870 are used to build the output buffer 1860 (e.g., aninverter) used to drive a signal onto the output 1880, which will havesome output load (not shown). Depending on the desired output, theoutput buffer 1860 could be replaced by more complex logic gates (e.g.,NAND or NOR) to build more functionality into the sense amplifier latchcircuit 1875 at minimal cost.

Clock inputs 1821, 1846, and 1831 are shown. Clock input 1831 is shownin figures above as being a SET line. Clock inputs 1821 and 1831 arecontrol inputs and are connected to the reference clock, φ. Clock input1846 is a control input and is connected to the opposite phase of thereference clock, φ.

Voltage waveforms that describe an exemplary operation of the senseamplifier latch circuit 1875 from FIG. 18 are shown in FIG. 19. As shownin FIG. 19 (with appropriate reference to FIG. 18), when the referenceclock, φ, is low, the precharge device 1820 is used to pull the inputnode 1811 to V_(DD). During this time, the opposite phase of thereference clock, φ, keeps the pass-gate device 1846 off to allow forprecharging of the input node 1811. The internal storage node 1801 canbe initialized to V_(DD) by passing through the precharge voltage from aprevious clock cycle. The source (e.g., and drain, if the source anddrain are coupled together) of the gated diode device 1831 is low (asthe clock input 831 is tied to φ), which places the gated diode 1830 inthe low capacitance state (e.g., only parasitic capacitances exist)because the channel in the gated diode 1830 is not inverted.

When the reference clock rises to the high state, the precharge device1820 turns off while the pass-gate 1845 turns on. If the signal on input1810 pulls the voltage on the input 1810 down toward zero potential, theinternal storage node 1801 voltage of the sense amplifier 1800 willbegin to drop. Because the source of the gated diode 1830 is now at ahigh voltage, the channel of the gated diode 1830 will become invertedas soon as the voltage at the internal storage node 1801 drops belowV_(DD)−|V_(Tp,gd)|, where V_(Tp,gd) is the threshold voltage of thegated diode 1830. This places the gated diode 1830 in a high capacitancestate. If the signal on input 1810 remains at V_(DD), the internalstorage node 1801 voltage will similarly remain at V_(DD), and the gateddiode 1830 will remain in the low capacitance state.

When the reference clock falls again to the low state, the source of thegated diode 1830 will also fall to zero potential. The pass-gate 1845 isturned off, which isolates the internal storage node 1801 of the senseamplifier 1800 such that no charge can be brought into or out of thisnode 1801. During this switching event, capacitive coupling between thetwo terminals (e.g., the gate coupled to internal storage node 1801 andthe source coupled to the clock input 1831) of the gated diode 1830 willresult in a drop in the potential of the internal storage node 1801. Ifthe gated diode 1830 was in the high capacitance state (i.e., dataduring the previous clock cycle was low), charge neutrality dictatesthat the internal storage node 1801 voltage will drop by a large amount.The exact value may be determined by the relative capacitance of thegated diode 1830 to the total capacitance of the internal storage node1801. This change in potential can be designed to switch the outputbuffer 1860 (if the voltage at the internal storage node 1801 fallsbelow V_(m,inv), the switching point of the buffer 1860), and afull-swing output voltage is obtained as the voltage on the output 1880switches to V_(DD).

If the gated diode 1830 was in the low capacitance state (i.e., dataduring the previous cycle of the reference clock was high), capacitivecoupling will be small—due only to parasitic elements—and the internalstorage node 1801 voltage will stay close to V_(DD) (the voltage fallsby only a small amount, ΔV_(par)) when the source of the gated diode1830 is pulled low. The output buffer 1860 therefore does not switch andan output voltage of zero is maintained on the output 1880. As such,this sense amplifier latch circuit 1875 outputs the logical complementof the input logic state. The input logic state can always bereconstructed by feeding the output 1880 of the buffer 1860 into anadditional inverting stage.

While the gated diode 1830 is evaluating the input data from theprevious high voltage of the reference clock, the first part 1802 (e.g.,the part of the circuit 1875 up to the pass gate 1845 and including theprecharge device 1820) of the circuit 1875 is again in the prechargestate since φ=0. When the reference clock switches high again, the firstpart 1802 of the circuit evaluates while the second part 1803 (e.g., thepart of the circuit from the pass gate 1845 through the output 1880) ofthe circuit 1875 performs an effective “precharge” operation by bringingthe voltage of the internal storage node 1801 back to a value largeenough to switch the output buffer 1860. Since there is likely a largecapacitance on the input node 1811, charge sharing would allow thepass-gate 1845 to bring the potential of the internal storage node 1801back to nearly VDD. As a result, a pipelined architecture is achievedwithin the sense amplifier latch circuit 1875 as the two adjacent parts1802 and 1803 of the circuit 1875 can simultaneously be in precharge andevaluate periods and vice versa. A similar sense amplifier latch circuitoperated with the opposite clock phase (e.g., replacing φ with φ and φwith φ) can be placed in parallel with this sense amplifier latchcircuit 1875, thus allowing data to be sensed and obtained during bothphases of the reference clock, thereby effectively doubling the datarate if the outputs of the two latches are multiplexed together. Datacan thus be transferred at twice the reference clock frequency.

For illustration purposes, the circuit 1875 in FIG. 18 is drawn usingPFETs for the precharge device 1820, pass-gate 1845, and gated diode1830. However, it is straightforward to design this circuit 1875 usinginstead NFET precharge devices 1820, pass-gates 1845, and gated diodes1830. In such a case, the polarity of the clock signals would bereversed.

In the circuit diagram shown in FIG. 18, both phases of the referenceclock are required (clock, φ, and its complement, φ). It mightdesirable, however, to use a single-phase clock design (φ only), thuseliminating the need to route two separate clock phases. This ensuresthat there are no timing problems due to skew between the two clockphases. Two such variations of the circuit shown in FIG. 18 arepresented in FIGS. 20 and 21.

FIG. 20 shows an example of a sense amplifier latch circuit 2075. Senseamplifier latch circuit 2075 comprises a precharge device 2020, a gateddiode amplifier 2000, an output buffer 2060, and keeper 2025. The gateddiode amplifier 2000 comprises a pass-gate 2045 and a gated diode 2030.The output buffer 2060 comprises a PFET 2065 and an NFET 2070. The senseamplifier latch circuit 2075 is coupled to an input 2010 and produces anoutput 2080. RC delay element 2015 is an exemplary element indicatingthat the input 2010 is a slow moving signal that needs amplification.Clock inputs 2021, 2031, and 2046 are control inputs and are coupled tothe reference clock, φ, which will typically be coupled to controlcircuitry, as described above.

In FIG. 20, an NFET pass-gate 2045 is used to isolate the internalstorage node 2001 from the input 2010. Because an NFET is used, thereference clock can be directly tied to the gate of the pass-gate 2045without inversion. However, since an NFET might not be able to pull theinternal storage node 2001 all the way up to the power supply voltage,V_(DD), a PFET keeper 2025 would likely need to be used. The PFET keeper2025 has a control input 2026 coupled to the output 2080.

In FIG. 21, the sense amplifier latch circuit 2175 comprises a prechargedevice 2120, a gated diode amplifier 2100, and an output buffer 2160.The gated diode amplifier 2100 comprises a pass-gate 2145 and a gateddiode 2130. The output buffer 2160 comprises a PFET 2165 and an NFET2170. The sense amplifier latch circuit 2175 is coupled to an input 2110and produces an output 2180. RC delay element 2115 is an exemplaryelement indicating that the input 2110 is a slow moving signal thatneeds amplification. Clock inputs 2121 and 2131 are control inputs andare coupled to the reference clock, φ, which will typically be coupledto control circuitry as described above. The control input 2146 for thepass-gate 2145 is coupled to the output 2180.

In FIG. 21, the PFET pass-gate 2145 is retained, but the control signal2146 applied to the gate of the PFET pass-gate 2145 is the output 2180.The only situation in which the pass-gate 2145 is switched off is whenthe voltage on the output 2180 changes (e.g., the gated diode 2130 pullsthe internal storage node 2101 low). This allows for isolation of theinternal storage node 2101 from the input 2110, which is prechargingduring the evaluation of the gated diode amplifier 2100. In addition,when the internal storage node 2101 is high, the pass-gate 2145 isturned on, which helps to stabilize the value of the internal storagenode 2101.

The circuit 2175 depicted in FIG. 21 is attractive, as compared to thecircuit 2075 of FIG. 20, because the circuit 2175 minimizes the numberof transistors needed to build the circuit 2175. Furthermore, thecircuit 2175 does not require that the input 2110 and pass-gate 2145operate to pull down the internal storage node 2101 (e.g., or 2001) asthe keeper 2025 (see FIG. 20) attempts to pull up the internal storagenode 2101 (e.g., or 2001), as would occur if the keeper 2025 were used.This contention for the internal storage node 2010 (e.g., or 2001) couldincrease delay and power dissipation.

However, when power is initially applied to the circuit 2175, it ispossible that the circuit 2175 could be stuck in a state from which thecircuit 2175 cannot recover (which is not true of the circuit 2075 ofFIG. 20). For example, if the internal storage node 2101 starts off atzero voltage and the clock at VDD, the output will never drop to zero.Capacitive coupling from the gated diode 2130 will simply drive theinternal storage node 2101 to a negative voltage, thus never flippingthe output buffer 2160. Thus, the pass-gate 2145 will never turn on, andthe internal storage node 2101 will never get precharged. This problemmay very well be mitigated by subthreshold or gate leakage currents thatmay charge up the internal storage node 2101 over time, but to ensurethat this problem is completely eliminated, circuits as shown in FIGS.22 and 23 may be used.

In FIG. 22, the sense amplifier latch circuit 2275 comprises a prechargedevice 2220, a gated diode amplifier 2200, and an output buffer 2260.The gated diode amplifier 2200 comprises a pass-gate 2245 and a gateddiode 2230. The output buffer 2260 comprises a PFET 2265 and an NFET2270. The sense amplifier latch circuit 2275 is coupled to an input 2210and produces an output 2280. RC delay element 2215 is an exemplaryelement indicating that the input 2210 is a slow moving signal thatneeds amplification. Clock inputs 2221 and 2231 are control inputs andare coupled to the reference clock, φ, which is typically coupled tocontrol circuitry as described above. One control input 2246 for thepass-gate 2245 is coupled to the output 2280 and another control input2247 for the pass-gate 2245 is coupled to the reference clock.

FIG. 22 thus adds a pass-gate 2245 made of an NFET portion 2248 inparallel with a PFET portion 2249—effectively combining the two circuitsshown in FIGS. 20 and 21. The NFET portion 2248 is useful in ensuringinitialization, as the NFET portion 2248 will allow for precharging ofthe internal storage node. Due to the threshold voltage drop required toturn on the NFET portion 2248 of the pass-gate 2245, the NFET portion2248 does not contribute significantly to the switching performance ofthe latch.

In FIG. 23, the sense amplifier latch circuit 2375 comprises a prechargedevice 2320, a gated diode amplifier 2300, an output buffer 2360, and aMOS protection diode 2350. The gated diode amplifier 2300 comprises apass-gate 2345 and a gated diode 2330. The output buffer 2360 comprisesa PFET 2365 and an NFET 2370. The sense amplifier latch circuit 2375 iscoupled to an input 2310 and produces an output 2380. RC delay element2315 is an exemplary element indicating that the input 2310 is a slowmoving signal that needs amplification. Clock inputs 2321 and 2331 arecontrol inputs and are coupled to the reference clock, φ, which istypically coupled to control circuitry as described above. A controlinput 2346 for the pass-gate 2345 is coupled to the output 2380.

FIG. 23 adds a MOS protection diode 2350 that prevents the internalstorage node 2301 from reaching a significant negative potential bysupplying charge to the internal storage node 2301. The charge ensuresthat when the reference clock switches high, capacitive coupling willpush the internal storage node 2301 far enough to switch the voltage onthe output 2380 to low, thus turning on the pass-gate 2345 and allowingthe precharge voltage to reach the internal storage node 2301.

Improved Gated Diode Structure for Low Vt, Low Vt Fluctuation and LowParasitic Capacitance

As CMOS technologies have scaled, the impact of manufacturing processvariations on device threshold voltages has become magnified. This isespecially important in conventional latch and sense amplifier circuits,which often rely upon cross-coupled inverters with matched thresholdvoltages for the switches making up the cross-coupled inverters. Thesense amplifier circuit and sense amplifier latch circuits proposedherein are dynamic circuits based upon charge storage and senseamplifiers based upon a gated diode, so that precise threshold voltagecontrol and matching is typically not required for proper circuitoperation. In addition, the critical devices in these sense amplifiercircuits and sense amplifier latch circuits can be designed to minimizethreshold voltage fluctuation without jeopardizing performance.

In the sense amplifier circuits and sense amplifier latch circuitsdescribed above, the switching or latching point for a small signalapplied at the input is determined at least by the threshold voltage ofthe gated diode (e.g., V_(T,gd)). While the sense amplifier circuits andsense amplifier latch circuits will still function if the thresholdvoltage of the gated diode parameter fluctuates, it is still desirableto minimize variation. Since the gated diode is not used to providecurrent drive, a large channel length, L, can be used. Thissubstantially eliminates threshold voltage variation due to gate length,critical dimension control, and the short-channel effect, as long as thelength is not too large such that carrier transit time across thechannel becomes significant. Similarly, channel width critical dimensioncontrol and the narrow-width effect can be avoided by designing a largedevice width, W. Both of these are also desirable to obtain a largecapacitance on the gated diode to maximize capacitive coupling to theinternal storage node during sensing. Furthermore, a low value ofV_(T,gd) is desirable because the voltage change, at the input, that isto be sensed will likely be small. This means that low dose ionimplantation will likely be used for the gated diode, which results in alower background doping concentration, and thus reduced susceptibilityto random dopant fluctuation and gate oxide thickness variation effectson the threshold voltage. All of these effects combined, the thresholdvoltage for the gated diode can be very precisely controlled.

In the simplest implementation of the sense amplifier circuits and senseamplifier latch circuits, the FET portion of the gated diode can be thesame as that of the MOSFETs. However, depending on the fabricationprocess flow, this may lead to excessive parasitic capacitance. Thus,while the gated diode itself is in the low capacitance state, anon-negligible capacitance exists that may couple to the internalstorage node of the sense amplifier. Much of this undesirablecapacitance arises from the gate overlap of the source and drain regionsof the device. In a conventional transistor, the size of this overlapregion must be carefully tuned to appropriately balance parasiticresistance and capacitance. This process normally results in the use ofan extension implant for the source and drain regions, which causessignificant overlap capacitance to the gate electrode. For example, atypical MOSFET 2400 is shown in FIG. 24. The MOSFET 2400 comprises deepsource/drain regions 2420, source/drain extensions 2430, and halos 2440.The source/drain extensions 2430 are added, as are the halos 2440, tobalance parasitic resistance and capacitance.

While optimal design of the gated diode must also balance parasitics, adifferent design point can be used since parasitic capacitance is moreimportant than parasitic resistance. Current drive, which is degraded byparasitic resistance, of the gated diode is only needed to charge anddischarge the series combination of the gated diode oxide capacitanceand any additional parasitics (e.g., anything tied to internal storagenode 1801 in FIG. 18), which can be significantly smaller than the fullgated diode oxide capacitance.

An improved design point that can be integrated into a standard processfabrication sequence is to simply mask out the extension and haloimplants. Removal of the extension implants greatly reduces parasiticcapacitance. The halo implants, which are normally used to controlshort-channel effects and the device threshold voltage, can also beremoved from the gated diode device because it is generally large inchannel length and of low threshold voltage. In addition, without thehalo implant, lateral encroachment of the source and drain regions isenhanced, which can reduce the series resistance penalty introduced byremoval of the extension implant.

FIG. 25 shows a capacitance versus gate voltage for the MOSFET of FIG.24. As can be seen, the capacitance at low gate voltages is relativelyhigh as compared to the capacitance (see FIG. 26) of a MOSFET formedwithout source/drain extensions 2430 and halos 2440. Thus, a MOSFETformed without source/drain extensions 2430 and halos 2440 can bebeneficial for a gated diode in the sense amplifier circuit and senseamplifier latch circuits described above.

It is to be understood that the embodiments and variations shown anddescribed herein are merely illustrative of the principles of thisinvention and that various modifications may be implemented by thoseskilled in the art without departing from the scope and spirit of theinvention.

1. A latch circuit comprising: a pass device comprising a controlterminal and first and second terminals, the first terminal of the passdevice coupled to a signal line, the control terminal of the pass devicecoupled to a first clock line; a precharge device comprising a controlterminal and first and second terminal, the control terminal of theprecharge device coupled to a second clock line, the first terminal ofthe precharge device coupled to a power supply voltage, and the secondterminal of the precharge device coupled to the first terminal of thepass device; a gated diode comprising first and second terminals, thefirst terminal of the gated diode coupled to the second terminal of thepass device, and the second terminal of the gated diode coupled to athird clock line; and an output device comprising an input and anoutput, the input of output device coupled to the first terminal of thegated diode and to the second terminal of the pass device, the output ofthe output device adapted to be the output of the latch circuit, theoutput device adapted to produce an output signal on the output of asense amplifier circuit based on a voltage on the first terminal of thegated diode; wherein the signal line is adapted to be coupled to aninput signal.
 2. The latch of claim 1, wherein the gated diode comprisesa Metal Oxide Semiconductor Field Effect Transistor (MOSFET), andwherein the MOSFET is formed with at least one deep source/drain regionbut without any source/drain extensions or halos.
 3. The latch circuitof claim 1, wherein the second and third clock lines are adapted to becoupled to a clock signal and wherein the first clock line is adapted tobe coupled to a complemented version of the clock signal.
 4. The latchcircuit of claim 3, the wherein precharge device and the pass deviceeach comprises a P-type Field Effect Transistor (PFET) and wherein thegated diode comprises a P-type device.
 5. The latch circuit of claim 1,wherein the output device comprises an inverter comprising a P-typeField Effect Transistor (PFET) and an N-type Field Effect Transistor(NFET).
 6. The latch circuit of claim 5, wherein: the PFET comprises acontrol terminal and first and second terminals, the control terminal ofthe PFET coupled to the first terminal of the gated diode, the firstterminal of the PFET coupled to a power supply voltage, and the secondterminal of the PFET coupled to a first terminal of the NFET; and theNFET comprises a control terminal, the first terminal and a secondterminal, the control terminal of the NFET coupled to the first terminalof the gated diode, and the second terminal of the NFET coupled atground; wherein the first terminal of the NFET comprises the output ofthe output device.
 7. The latch circuit of claim 1, wherein the first,second and third clock lines are adapted to be coupled to a single clocksignal.
 8. The latch circuit of claim 7, wherein precharge devicecomprises a P-type Field Effect Transistor (PFET), wherein the passdevice comprises an N-Type FET (NFET) and wherein the gated diodecomprises a P-type device.
 9. The latch circuit of claim 1, furthercomprising a keeper device comprising a control terminal and first andsecond terminals, the control terminal of the keeper device coupled tothe output of the latch circuit, the first terminal of the keeper devicecoupled to the voltage supply, and the second terminal of the keeperdevice coupled to the first terminal of the gated diode.
 10. The latchcircuit of claim 9, wherein the keeper comprises a P-type Field EffectTransistor (PFET).
 11. The latch circuit of claim 1, wherein the firstclock line coupling the output of the latch circuit with the controlterminal of the pass device and wherein the pass device comprises aP-type Field Effect Transistor (PFET).
 12. The latch circuit of claim11, wherein the latch circuit further comprises a protection diodehaving a control terminal and an output terminal, output device iscoupled to the voltage source and to ground, the control terminal of theprotection diode is coupled to ground and the output terminal of theprotection diode is coupled to the first terminal of the gated diode.13. The latch circuit of claim 12, wherein the protection diodecomprises an N-type FET (NFET) comprising the control terminal and firstand second terminals, the first terminal of the NFET of the protectiondiode being the output terminal of the protection diode, and the secondterminal of the NFET of the protection diode coupled to the controlterminal of the NFET of the protection diode.
 14. The latch circuit ofclaim 1, wherein the first clock line comprises a line coupling theoutput of the latch circuit with the control terminal of the passdevice, the pass device comprises a P-type Field Effect Transistor(PFET), the pass device further comprises an N-type Field EffectTransistor (NFET) in parallel with the PFET, the NFET comprising acontrol terminal and first and second terminal, the control terminal ofthe NFET coupled to the third clock line, the first terminal of the NFETcoupled to the signal line, and the second terminal of the NFET coupledto the first terminal of the gated diode.
 15. A semiconductor comprisingat least one latch circuit, each of the at least one latch circuitscomprising: a pass device comprising a control terminal and first andsecond terminal, the first terminal of the pass device coupled to asignal line, the control terminal of the pass device coupled to a firstclock line; a precharge device coupled to a power supply voltage, andthe second terminal of the precharge the control terminal of theprecharge device coupled to a second clock line, the first terminal ofthe precharge device coupled to a power supply voltage, and the secondterminal of the precharge device couple to the first terminal of thepass device; a gated diode comprising first and second terminals, thefirst terminal of the gated diode coupled to the second terminal of thepass device, and the terminal of the gated diode coupled to a thirdclock line; and an output device comprising an input and an output, theinput of the output device coupled to the first terminal of the gateddiode and to the second terminal of the pass device, the output of theoutput device adapted to be the output of the latch circuit, the outputdevice adapted to produce an output signal on the output of the senseamplifier circuit based on a voltage on the first terminal of the gateddiode; wherein the signal line is adapted to be coupled to an inputsignal.